An experimental study of an antenna system for a portable terminal

May 26, 2017 | Autor: Ammar Sharaiha | Categoria: Experimental Study, Optical physics, Electrical And Electronic Engineering
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Electromagnetic Wa¨ es in Chiral and Bi-isotropic Media, Artech, Boston, 1994. I. V. Lindell and Weiglhofer, ‘‘Green Dyadic and Dipole Fields for a Medium with Anisotropic Chirality,’’ IEE Pt. H, Vol. 141, 1994, pp. 211]215. K. Hayata and M. Koshiba, ‘‘Chirosolitons: Unique Spatial Solitons in Chiral Media,’’ IEEE Trans. Microwa¨ e Theory Tech., Vol. MTT-43, 1995, pp. 1814]1818. I. V. Lindell, A. H. Sihvola, P. Puska, and L. H. Ruotanen, ‘‘Conditions for the Parameter Dyadics of Lossless Bianisotropic Media,’’ Microwa¨ e Opt. Technol. Lett., Vol. 8, 1995, pp. 268]272. D. Cheng, On the Electromagnetic Field Theory in Quasi-Chiral Materials, Ph.D. dissertation, University of Electronic Science and Technology of China, 1995.

Q 1997 John Wiley & Sons, Inc. CCC 0895-2477r97

AN EXPERIMENTAL STUDY OF AN ANTENNA SYSTEM FOR A PORTABLE TERMINAL M. Clenet,1 J. F. Diouris,1 A. Sharaiha,2 J. Saillard,1 and C. Terret 2 1 Laboratoire Systemes Electroniques et Informatiques ` EP CNRS 63 I.R.E.S.T.E. ATLANPOLE, La Chantrerie, 44 087 NANTES, France 2 Laboratoire Structures Rayonnantes U.R.A au CNRS No. 834 Universite ` de Rennes 1 Avenue du General ´ ´ Leclerc 35 042 Rennes Cedex, France Recei¨ ed 7 No¨ ember 1996 ABSTRACT: The subject is the study of an antenna system for a portable terminal. Experiments and theoretical results of the different types of antennas we use are compared. The main characteristics of the antenna system are reported and compared. A blind adapti¨ e algorithm is used with the experimental data in order to impro¨ e reception by using space di¨ ersity. Q 1997 John Wiley & Sons, Inc. Microwave Opt Technol Lett 14: 354]360, 1997. Key words: wire antenna; printed antenna; space di¨ ersity; multisensor processing 1. INTRODUCTION

In the last few years, much interest has been generated in portable telephones and data terminals inside towns and buildings. In a radio environment with noise caused by fading effects and intersymbol interference caused by multipath propagation, the use of space diversity is required to improve reception. In addition new possibilities in the miniaturization of high-frequency components allow us to consider the use of several sensors on a cellular telephone. Therefore, we have studied a four-sensor portable terminal: two wire antennas on the top of the handset, and two stacked printed antennas on each side. The purpose of this article is to describe this antenna system, and to show the multisensor reception improvement by using measurements. The article is organized as follows: v

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First the two different types of antennas used for our system are described.

v

v

v

Then the handset and the measurement system we have designed are presented. The result of the characterization of our antenna system, computed with data collected in an anechoic chamber, is reported. Finally, the improvement obtained with the use of a blind adaptive multisensor algorithm with measured data is presented.

2. THE ANTENNA SYSTEM

An indoor or urban environment is characterized by a multipath propagation. The recombination of the different paths can be constructive at one position, and destructive a few centimeters away. This phenomenon is known as fast fading. Space diversity can help to fight against this effect. But for an optimum result, each antenna must have a large half-power beamwidth, because the handset user does not always point out in the transmitting direction. Thus, we need antennas that have a good space coverage for vertically and horizontally polarized waves. In order to eliminate the losses due to a matching network, these antennas must also have a good matching to 50 V. Our choice was to use four antennas: two wire antennas placed on top of the handset, and two printed antennas placed on each side ŽFigure 1.. The two wire antennas are monopole antennas loaded with a modified folded dipole. Their geometry and dimensions as functions of the wavelength are shown in Figure 2. This kind of antenna was previously proposed by Altshuler w1x. The dimensions have been optimized in order to improve the bandwidth and the omnidirectional performance. For theoretical analysis, a numerical electromagnetic code, obtained from integral equations for wire antennas, has been used. Results. For the geometrical dimensions given in Figure 2, a match for a voltage standing wave ratio ŽVSWR. lower than 2 is obtained in a 6.5% bandwidth. The computed radiation patterns of the monopole loaded with a folded dipole, in the two main planes in vertical and horizontal polarization are compared to the radiation patterns of a simple quarter-wavelength monopole in Figure 3. The antennas are over an infinite ground plane. In the plane w s 08 Žthe orthogonal plane to the folded dipole., a reception improvement is obtained for vertically polarized waves compared to the radiation pattern of a simple monopole, especially in the axis where the null has disappeared. In the plane w s 908 Žthe plane that contains the folded dipole., the loaded monopole can receive horizontally polarized waves. These two phenomena are due to the contribution of the folded dipole. But over an infinite ground plane, the radiation pattern in the plane w s 908 in vertical polarization is the same for the loaded monopole and for the simple monopole: There is always a null in the axis. Figure 4 presents the computed radiation patterns of the monopole loaded with a folded dipole in the two main planes and for the two orthogonal polarizations when it is placed at a distance of l 0r8 from the center of the top of the 0.45l0 = 0.35l0 = 1.53 l0 conducting box. In the plane w s 08, this antenna has a good coverage of space in vertical polarization. But some hollows appear in the direction u s y1208 and u s y1658, for example. They are due to the finite ground plane, and also to the fact that this antenna is not symmetric. In the plane w s 908, a good coverage of the space is ob-

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synthesizer is used as a local oscillator. The local oscillator is common to each channel in order to achieve coherent processing. The four-channel power divider is designed in microstrip technology with three Wilkinson dividers. Each rf microstrip line is very close to the others because the volume available is small. So, there is a significant electronic coupling between the channels Ž17 dB.. Figure 1

Figure 2

Design of the four-sensor portable terminal

Geometry and dimensions of the loaded monopole

4. THE MEASUREMENT SYSTEM

The measurement system is described in Figure 9. The transmitter consists essentially of a signal generator, a frequency converter, and a power amplifier. The transmitting antenna is a monopole antenna loaded with one folded dipole. The signal generator creates a random sequence of variable length, and realizes a differential QPSK modulation with a 2-Mbitrs transmission rate. The acquisition system consists of our portable receiver connected to a four-channel digital oscilloscope, which digitizes the received signal at the frequency of 25 MHz, and a computer that performs a Hilbert filtering

tained. The null in the axis has disappeared, which is not the case when this antenna is over an infinite ground plane. This is due to the fact that the antenna is not in the center of the top of the conducting box. The radiation patterns in horizontal polarization indicate the good coverage in the two main planes. Finally, notice that almost all the variations in the space of the radiated power in the two orthogonal polarizations are less than 10 dB. Figure 5 presents the measured radiation pattern in the plane w s 08 of the wire antenna when it is placed on the left side of the top of the conducting box. There is a good agreement with the theoretical results. The mean maximum gain is around 6.4 dB in the bandwidth. The undulations around u s 908 are due to the presence of the conducting box and the right wire antenna between the incoming wave and the measured wire antenna. The microstrip antennas are composed of two substrate layers in order to improve the bandwidth w2x. The lower layer is a high-permittivity material, which is used in order to reduce the dimensions. Their geometry and their dimensions as functions of the wavelength are shown in Figure 6. The lower patch is fed on the diagonal; thus this antenna can receive vertically and horizontally polarized waves in the two main planes. Figure 7 presents the radiation pattern in the plane w s 08 of the microstrip antenna when it is placed on the left side of the conducting box. It shows that this antenna can receive vertically and horizontally polarized waves in the plane that contains the four antennas. The half-power beamwidth is 908 in vertical polarization and 1108 in horizontal polarization. The mean maximum gain is around 3.5 dB in the bandwidth. The electromagnetic coupling has also been measured. The maximum is obtained between the two wire antennas and reaches a value of y15.4 dB. 3. CONSTITUTION OF THE HANDSET

A four-sensor portable terminal prototype has been designed. Two wire antennas are placed on top of the handset and two stacked printed antennas are put on each side, as shown in Figure 1. Inside the conducting box, there are four identical processing channels ŽFigure 8.. Each consists of a low-noise amplifier, a mixer that converts the rf signal at 6.5 MHz i.f. frequency, an i.f. amplifier, and a bandpass filter. A frequency

Figure 3 Radiation patterns of the loaded monopole Žcrosses. and the simple monopole Žcircles.. Ža. Plane w s 08, vertical polarization; Žb. plane w s 908, vertical polarization; Žc. plane w s 08, horizontal polarization

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Figure 4 Radiation patterns of the loaded monopole Žcrosses. and a simple monopole Žcircles . placed at a lr8 distance from the center of the conducting box. Ža. Plane w s 08, vertical polarization; Žb. plane w s 908, vertical polarization; Žc. plane w s 08, horizontal polarization; Žd. plane w s 908, horizontal polarization

Figure 6

Geometry and dimensions of the stacked printed antenna

in order to obtain phase and quadrature projections of the signal. The computer also records data, for delayed processing. 5. CHARACTERIZATION OF THE ANTENNA SYSTEM

Figure 5 Measured radiation pattern of the loaded monopole placed at a lr8 distance from the center of the conducting box

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The first measurement campaign was made in an anechoic chamber, to evaluate the source vector with an experimental receiver. The knowledge of the source vector helps to estimate the performance of the processing of the signal. The measurements were made for an azimuth angle Ž w . varying

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Figure 8

Electronic internal postprocessing

Figure 7 Measured radiation pattern of the printed antenna on the conducting box

from 08 to 1708 in 108 steps, and an elevation angle Ž u . varying from y1208 to 1208 in 108 steps. For example, Figure 10 shows the amplitude of the received signal in two directions on the four antennas for two orthogonal polarized waves. Note the difference in level and phase of the received signal between the four sensors, and between two different angles of the incoming wave. After having estimated the source vector for all the directions, an en¨ elope correlation coefficient was computed with the use of spatial samples. The envelope correlation coefficient indicates the performance of diversity processing. The smaller its value, the better the diversity processing will be. The results are given in Table 1. They show that the four channels are partly correlated. This is because the distance between antennas is too short, and there is an electromagnetic and an electronic coupling inside the handset. The ability of an array to steer one beam and one or more nulls simultaneously is determined by several factors, such as the element position, the element antenna radiation pattern, the polarization of the signals, and the direction of the beam and the nulls. The spatial correlation coefficient b , introduced by Lin w4x, includes these factors and completely characterizes array beam pointing and nulling. Figure 11 shows the spatial correlation coefficient of our system versus the standard deviation in elevation between two directions of arrival in the two main planes. b decreases with the increase of the difference of incidence of the two incoming waves. It can be noticed that b is worse in the plane w s 908 than in the plane w s 08, because the plane w s 08 contains the four antennas and so has more degrees of freedom. Second, some measurements were carried out in an electronics laboratory. In this case, the propagation is characterized by multipath. The transmitter generates a QPSK modulated signal with a 2-Mbitrs rate. The transmitter and the receiver were placed 1.2 m above the floor, and they were 4 m apart. Figure 12 shows an example of the received signal on each antenna after frequency recovery. The signal-to-noise ratio ŽSNR. is computed in the middle of each symbol, after phase recovery. The SNR is quite different on the four antennas, due to the presence of fast fading. Several acquisitions were made for different positions of the handset. In Figure 13Ža., the measured voltage on sensors 2]4 versus the measured voltage on sensor 1 are plotted. A

cloud of points is obtained; this shows that the received signal on each sensor is little correlated when there is a multipath propagation. In Figure 13Žb., the estimated signal-to-noise ratio of sensors 2]4 versus the SNR of sensor 1 are indicated. This plot confirms that the four signals are little correlated. A previously studied w5x blind adaptive multisensor algorithm ŽFigure 14. was used in order to evaluate the multisensor improvement qualitatively. The QPSK coherent demodulator realizes the carrier phase recovery by using a costa loop, and the symbol timing recovery by using an early-late synchronizer. The weight w k are calculated by mean-square error criterion with the use of the recursive least-squares algorithm. Figures 15Ža. and 15Žb. presents the eye diagrams of the received signal on sensors 1 and 3, and the corresponding SNRs are mentioned. Figure 15Žc. presents the eye diagram of the signal at the output of the adaptive receiver. The eye diagram is more open, and the SNR at the output equals 20 dB. There is a good agreement between the experimental result and the theoretical result obtained in the case of a frequency nonselective channel Ž20.4 dB.. 6. CONCLUSION

We have designed a multisensor receiver prototype for indoor and urban communications. The antenna elements have a good space coverage. They also have good matching over a large bandwidth. Many measurement campaigns have been carried out in anechoic chambers as well as in indoor environments. The characterization of our antenna system has

Figure 9

Constitution of the measurement system

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Figure 10 Amplitude of the received signals of the four sensors in the two polarizations. Solid lines, antenna 1; dashed lines, antenna 2; dot-dashed lines, antenna 3; dotted lines, antenna 4. Ža. w s 08, u s 508, vertical polarization; Žb. w s 08, u s 508, horizontal polarization; Žc. w s 08, u s y1008, vertical polarization; Žd. w s 08, u s y1008, horizontal polarization TABLE 1

Envelope Correlation Coefficient 1

2

3

4

Antenna

1

0.28 1

0.26 0.45 1

0.45 0.41 0.44 1

1 2 3 4

Figure 11

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Spatial correlation coefficient. Ža. Plane w s 08, Žb. plane w s 908

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Figure 12

Figure 13 sensor

Example of measured signals on each sensor after carrier recovery

Ža. Comparison of the measured voltage on each sensor, Žb. comparison of the estimated signal-to-noise ratio on each

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‘‘Multisensor Receiver for Mobile Communications: An Experimental Study,’’ 28th Asilomar Conf. on Signals, Computers and Systems, 1994. Q 1997 John Wiley & Sons, Inc. CCC 0895-2477r97

Figure 14

Block diagram of the blind adaptive multisensor receiver

FORMULAS FOR THE COMPUTATION OF THE FAR-FIELD RADIATION PATTERNS OF RECTANGULAR MICROSTRIP ANTENNA ELEMENTS WITH THICK SUBSTRATES Mehmet Kara1 Weapons Systems Division Defence Science and Technology Organisation PO Box 1500, Salisbury, SA 5108, Australia

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Recei¨ ed 13 No¨ ember 1996 ABSTRACT: Formulas based on the two-slot, the ca¨ ity, and the surface current models to compute the far-field radiation patterns in both the E and H planes for rectangular microstrip antenna elements are studied and their ¨ alidity assessed. Three closed-form formulas are presented for the calculation of the radiation patterns in the E plane for antenna elements with substrates thicker than those reported elsewhere. These are deri¨ ed by modifying the two-slot model by the use of empirically deri¨ ed correction factors, taking into account the power radiated in space and surface wa¨ es, the surface-wa¨ e model coefficients, and the line extension. The capability of the de¨ eloped formulas is illustrated by comparison between the computed results and experimental measurements. The computed results are in good agreement with measurements. The H-plane patterns are computed from the two-slot model andror the ca¨ ity models. The half-power beamwidth in the E and H planes ha¨ e also been determined for each of the antenna elements in¨ estigated in this work. Q 1997 John Wiley & Sons, Inc. Microwave Opt Technol Lett 14: 360]367, 1997. Key words: radiation patterns; far-field patterns; microstrip antenna patterns; rectangular parch antennas 1. INTRODUCTION Figure 15 Eye diagrams of the received signal. Ža. On sensor 1, SNR s 11.5 dB; Žb. on sensor 3, SNR s 17.0 dB; Žc. at the output of the blind adaptive four-sensor receiver, SNR s 20.0 dB ŽSNR th s 20.4 dB.

shown that the capability of the handset to reject a jammer is not very good, but it is well suited for diversity reception. REFERENCES 1. E. E. Altshuler, ‘‘A Monopole Antenna Loaded with a Modified Folded Dipole,’’ IEEE Trans. Antennas Propagat., Vol. AP-41, No. 7, 1993, pp. 871]876. 2. A. B. Smolders and M. E. J. Jeuken, ‘‘Broadband Microstrip Phased-Array Antennas on a High-Permittivity Substrate,’’ JINA, Nice, France, 8-10 nov. 1994, pp. 650]653. 3. W. C. Y. Lee, Mobile Communications Engineering, MacGraw-Hill, New York, 1982. 4. H-C. Lin, ‘‘Spatial Correlations in Adaptive Arrays,’’ IEEE Trans. Antennas Propagat., Vol. AP-30, No. 2, 1982, pp. 212]223. 5. J. F. Diouris, J. Saillard, A. C. Tarot, C. Terret, and J. P. Blot,

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Radiation from a microstrip antenna occurs mainly from the fringing fields between the edge of the patch conductor and the ground plane, as shown in Figure 1. The fields at the edges can be resolved into normal and tangential components with respect to the ground plane. The tangential components Žthose parallel to the ground plane. are in phase, and are combined to give a maximum radiated field normal to the ground plane. Several methods have been proposed for the calculation of the far-field radiation patterns of microstrip antenna elements w1]15, 17]19x. Many of these analysis techniques employ approximations that are valid only for thin substrates. Other more general methods suffer from a lack of computational efficiency, which in practice can restrict their usefulness because of high computational time and costs. Based on this observation, this article investigates formulas based on the two-slot model ŽTSM., the cavity model ŽCM., and the electric surface current model ŽESCM. methods for calculating the far-field radiation patterns in both the E and the H planes, respectively. This is because of their applicability to the pattern analysis of rectangular microstrip

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