out this work. Special thanks are due to the engineers of Microwave Sensors Antenna Division, Antenna Systems Area, SAC for their help and necessary support. The authors also wish to thank Mr. Y. H. Trivedi, Mr. A. V. Apte, Mr. Rajesh Patel, Mr. Viren Sheth, Mr. H. S. Solanki, and Mr. K. P. Raja for providing necessary support in carrying out the antenna measurements. Dhaval Pujara would like to thank the management of Nirma University, Ahmedabad, for sponsoring him to SAC to carry out this research work. REFERENCES
Figure 8 Measured secondary radiation pattern of an offset reflector illuminated by a dual-mode corrugated horn (a) For Phi ¼ 0 and (b) For Phi ¼ 90
results validate the effectiveness of the dual-mode corrugated horn as a primary feed device for an offset reflector antenna. 4. CONCLUSION
In this article, the design of a novel dual-mode corrugated horn has been presented. Some remarks on the design and the results are in order: i. The dual-mode corrugated horn, can effectively suppress the unwanted high cross-polarization introduced by the offset geometry in an offset parabolic reflector antenna. ii. The amplitude proportion of the HE21 mode largely depends on the offset geometry. Generally, it is of the order from 15 dB to 30 dB with respect to the fundamental HE11 mode. iii. The HE21 mode should satisfy a quadrature phase (90 ) relationship with the fundamental (HE11) hybrid mode. iv. Utmost care should be taken while deciding the horn dimensions, and also during the fabrication process to ensure the desired improvement in the cross-polarization of the offset reflector. ACKNOWLEDGMENTS
The authors sincerely thank, the Director, Space Applications Centre (SAC), Ahmedabad, India for his encouragement to carry
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1. Y.T. Lo and S.W. Lee, Antenna handbook, Part B, Chapter 22, Van Nostrand Reinhold Company, New York. 2. W. Rudge and N.A. Adatia, Offset-parabolic reflector antennas: A review, Proc IEEE 66 (1978), 1592–1618. 3. J.L. Volakis, Antenna engineering handbook, The Mc-Graw Hills Companies, pp. 41.6–41.8. 4. T.-S. Chu and R.H. Turrin, Depolarization properties of offset reflector antennas, IEEE Trans Antennas Propag AP-21 (1973), 339–345. 5. J. Dijk, C.T.W. Van Diepenbeek, E.J. Maanders, and L.F.G. Thurlings, The polarization losses of offset paraboloid antennas, IEEE Trans Antennas Propag AP-22 (1974), 513–520. 6. J. Jacobsen, On the cross polarization of asymmetric reflector antenna for satellite applications, IEEE Trans Antennas Propag (1977), 276–283. 7. W. Strutzman and M. Terada, Design of offset-parabolic-reflector antennas for low cross-pol and low sidelobes, IEEE Antennas Propag Mag 35 (1993), 46–49. 8. A.D. Olver, et al., Microwave horns and feeds, Chapter 9, IEEE Press, 1994. 9. A.W. Rudge and N.A. Adatia, New class of primary-feed antennas for use with offset parabolic reflector antennas, Electron Lett 11 (1975), 597–599. 10. W. Rudge and N.A. Adatia, Matched-feeds for offset parabolic reflector antennas, In: Proceedings of 6th European Microwave Conference, Rome, Italy, 1976, pp. 143–147. 11. K. Watson, A.W. Rudge, and N.A. Adatia, Dual-polarized mode generator for cross-polar compensation in parabolic reflector antennas, In: Eighth European Microwave Conference, Paris, 1978, pp. 183–187. 12. P.J.B. Clarricoats and P.K. Saha, Propagation and radiation behavior of corrugated feeds, Proc IEE 118 (1971), pp. 1167–1176. 13. W. Rudge, Multiple-beam antennas: Offset reflectors with offset feeds, IEEE Trans Antennas Propag 23 (1975), 317–322. 14. C. Granet and G.L. James, Design of corrugated horns- a primer, IEEE Antennas Propag Mag, 47 (2005), pp. 76–84. C 2009 Wiley Periodicals, Inc. V
A 0.6-V LOW-POWER ARMSTRONG VCO IN 0.18 lM CMOS Cheng-Chen Liu, Sheng-Lyang Jang, Jhao-Jhang Chen, and Miin-Horng Juang Department of Electronic Engineering, National Taiwan University of Science and Technology, 43, Keelung Road, Section 4, Taipei, Taiwan 106, Republic of China; Corresponding author:
[email protected] Received 14 April 2009 ABSTRACT: A low-power differential voltage-controlled oscillator (VCO) is proposed and implemented in a 0.18 lm CMOS 1P6M process. It consists of two single-ended Armstrong oscillators via cross-coupled transistors to obtain a differential output. At the supply voltage of 0.6 V, the output phase noise of the differential VCO is 120.02 dBc/Hz at 1 MHz offset frequency from the carrier frequency of 3.85 GHz and the figure of merit is 188.5 dBc/Hz. Total VCO core
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DOI 10.1002/mop
power consumption is 2.1 mW. Tuning range is about 550 MHz, from 3.81 to 4.36 GHz, whereas the control voltage was tuned from 0 to C 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett 2.0 V. V 52: 116–119, 2010; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.24864 Key words: CMOS; differential Armstrong oscillator; VCO; low power; transformers
1. INTRODUCTION
Wireless communication rapidly develops to turn this technique into an important industry at present. The CMOS technology easily integrates the radio frequency (RF) circuit because it has the features of low power and lower cost. The VCO is a significant building block in modern RF transceiver, and it is used to generate the local signal to the up-mixer or down-mixer. Many studies have made efforts in the CMOS VCO design to improve the VCO performance. The classical Armstrong VCO [1] was invented in 1915, since then many integrated Armstrong VCOs have been developed [2–4]. Two configurations for the Armstrong VCOs have been adopted in the past: one is the balanced Armstrong VCO [5] where two single-ended Armstrong VCOs are coupled with balanced varactors; the second approach [3] uses two singleended Armstrong VCOs, the drain inductor in one VCO is coupled to the gate inductor in the other VCO and vice versa.
Figure 2 (a) A simplified half-circuit of the proposed Armstrong VCO without cross-couple transistors. (b) Simplified small-signal equivalent circuit
This article proposes a new Armstrong VCO with high performance, and the VCO consists of two single-ended Armstrong oscillators with a pair of cross-coupled transistors to generate differential output. The figure of merit (FOM) of this VCO is 188.5 dBc/Hz. The design of the proposed VCO circuit is presented in Section 2. Experimental results are shown in Section 3. Finally, the conclusion follows in the last section. 2. DESIGN OF DIFFERENTIAL ARMSTRONG VCO
Figure 1(a) shows the conventional differential Armstrong VCO, which includes two single-ended Armstrong oscillators composed of an nMOSFET Mn3 (Mn4) and the two inductors (L5, L3), the two inductors are configured as a transformer used for feedback and resonator’s inductor. The Mn3 (Mn4) is used to provide one negative differential resistance to compensate for the loss of LC-tank, which consists of varactors (CV) and inductors (L5). Figure 1(b) shows the proposed differential VCO circuit. The inductors (L1 and L2) form a center-tap inductor and provide a gate voltage to the Mn3 (Mn4). The capacitors (C1, C2) are used to improve the phase noise performance, and the varactors (CV) are used to tune the oscillation frequency. The main role of cross-coupled transistors (Mn1 and Mn2) is to force the two single-ended VCOs run in differential mode. Figure 2(a) shows the half-circuit topology of the proposed VCO without cross-coupled transistors (Mn1 and Mn2) in Figure 1(b). The single-ended Armstrong VCO’s small signal model is shown in Figure 2(b). The LT1 and LT2 are transformer’s input/ output coupling inductance, respectively. The CT includes input capacitor C1 of miller effect and parasitic capacitor Cgd (gatedrain) of Mn3, and its equivalent capacitor is as follows: CT ¼ ðC1 þ Cgd Þð1 AvÞ;
Figure 1 (a) Schematic of Armstrong VCO [4]. (b) Schematic of the proposed transformer coupled VCO
DOI 10.1002/mop
(1)
where Av is voltage gain from gate to drain of Mn3, and it is about gmZout. gm is transconductance and Zout is output impedance consisted of the C1, which is parallel with LT2. Cgs is the parasitic capacitor (gate-source) of Mn3. The input impedance Zin is calculated according to the small signal model shown in Figure 2(b) as follows
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Figure 5 Simulated, respectively, the L3 (L4)/L5 (L6) inductance, Qfactor, and coupling coefficient k of the transformer
Figure 3 Simulated the transient gate voltage of Mn2 (Vg) and drain voltage of Mn2 (Vd). VDD ¼ 0.6 V and VTUNE ¼ 0 V
Zin ¼
jxLT1 L1 : ðLT1 þ L1 Þ x2 LT1 L1 ðCT þ Cgs þ Cv Þ
(2)
When Zin ¼ 1, the parallel resonator generates a oscillation frequency xo, which angular frequency is equal to sffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi LT1 þ L1 : (3) xo ¼ LT1 L1 ðCT þ Cgs þ Cv Þ The oscillation frequency of the proposed VCO is different from [3] because of the presence of (Mn1 and Mn2), the parasitic capacitance which decreases the oscillation frequency. Figure 3 shows the simulated transient gate voltage (Vg) and drain voltage (Vd) of Mn2 in Figure 1(b). The second role of (L3, L4) is to boost the gate voltage swing of (Mn1 and Mn2). The gate voltage swing is larger than the drain voltage swing, so that the transistors (Mn1 and Mn2) can operate at lower supply voltage. The second role of (Mn1 and Mn2) is to generate an additional negative differential resistance to the resonator. This speeds up the start-up transient oscillation. Figure 4 shows the photograph of the fabricated VCO with chip area of 0.951 0.589 mm2 including all test pads and dummy metal. L3 and L5 (L4 and L6) in Figure 1(b) are laid out as one four-port transformer with five-turn inductors, which use metal layer six with 2.34 lm thickness and 9 lm width. The transformer design uses the ADS momentum tools, in which the simulated inductances/series resistances for the single-turn L3 and four-turn L5 (L4 and L6) are 0.48 nH/2.2 X and 3.3 nH/10.2
Figure 4
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Chip photograph of the proposed VCO
Figure 6
Measured tuning range of the VCO. VDD ¼ 0.6 V
X at 3.85 GHz, respectively. Figure 5 shows the transformer’s parameters including the L3 and L5 inductance, quality factor (Q), and coupling coefficient (k). The transformer was designed at the maximum Q-factor to improve VCO’s phase noise. 3. MEASUREMENT AND DISCUSSION
The proposed was designed and implemented in the TSMC 0.18 lm 1 P6M CMOS process. The VCO was measured with an Agilent E4407B spectrum analyzer. The circuit is biased at a 0.6-V supply voltage. Figure 6 shows the measured tuning curve
Figure 7 Measured output spectrum of the proposed VCO at 3.85 GHz. The output power is 13.33 dBm. VDD ¼ 0.6 V and VTUNE ¼ 0.2 V
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TABLE 2 Ref. [5] [7] [8] [9] This work
Performance Comparision With Previous Reports Freq (GHz)
VDD (V)
PN@1 MHz (dBc/Hz)
PDC (mW)
FOM (dBc/Hz)
5.12 3.6 4.56 4.5 3.85
0.6 1.8 1.8 1.5 0.6
116.7 126.8 121 122.47 120.02
3.9 10.8 7.56 6.75 2.1
185 187.6 185 187 188.5
Figure 8 Measured phase noise of the proposed VCO. VDD ¼ 0.6 V and VTUNE ¼ 0.2
by varying the varactor tuning voltage. As the tuning voltage VTUNE sweeps from 0 to 2 V, the varactor’s capacitance decreases, and the oscillation frequency varies from 3.81 to 4.36 GHz, indicating a tuning range of 550 MHz. The current and power consumption of the balanced VCO without buffers are 3.5 mA and 2.1 mW, respectively. Figure 7 shows the measured output spectrum at the center oscillation frequency of 3.85 GHz. The output power is 13.33 dBm. The measured phase noise is shown in Figure 8, and it is 120.02 dBc/Hz at 1 MHz offset frequency from the center frequency of 3.85 GHz. The FOM of the proposed VCO is about 188.5 dBc/Hz, which is calculated using the FOM defined as [6] x o FOM ¼ LfDxg þ 10 logðPDC Þ 20 log ; (4) Dx where L{Dx} is the SSB phase noise measured at Dx offset from xo carrier frequency and PDC is the DC power consumption in mW. Table 1 summarizes the circuit performance of the proposed VCO. The performance comparison between the proposed Armstrong CMOS VCO with other earlier similar frequency-band VCOs are summarized in Table 2. Figure 9 shows the phase noise and the output power variation as the control voltage was tuned from 0 to 0.6 Vat VDD ¼ 0.6 V. The data show that the VCO has a low-phase noise performance across the whole tuning range. 4. CONCLUSIONS
A novel differential Armstrong voltage-controlled oscillator (VCO) has been proposed and successfully implemented in the 0.18 lm CMOS process. The proposed integrated VCO circuit consists of two identical single-ended Armstrong VCOs with a pair of cross-coupled transistors to enhance the VCO performance so that the Armstrong VCO can operate at low voltage and low supply voltage. The proposed VCO provides differential outputs with the FOM 188.5 dBc/Hz. The power consumption of the VCO is 2.1 mW when the supply voltage is 0.6 V. TABLE 1
Performance Summary
Technology Supply voltage DC power (Core) Free-running Phase noise @1 MHz
Unit
Performance
– V mW GHz dBc/Hz
TSMC 0.18 lm CMOS 0.6 2.1 3.81–4.36 120.02
DOI 10.1002/mop
Figure 9 Measured phase noise at 1 MHz offset frequency is, respectively, vs. output power and turning voltage
ACKNOWLEDGMENTS
The authors thank the Staff of the National Chip Implementation Center (CIC) for the chip fabrication and technical supports.
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