A CMOS 0.35μm, 1.5V multi-band RF Q-enhanced LC bandpass filter

A CMOS 0.35µm, 1.5V Multi-band RF Q-enhanced LC Bandpass filter Aymen Ben Hammadi, Mongia Mhiri, Kamel Besbes Microelectronic and Instrumentation LR FSM, University of Monastir Monastir, Tunisia [email protected], [email protected], [email protected] Abstract— In this paper, a 2nd order tunable RF CMOS LC bandpass filter is presented. Q-enhancement and tuning are also considered. The Q-tuning is implemented through an adjustable negative conductance generator circuit. The simulated results show that using a 0.35 µm AMS CMOS process, the filter operates at center frequency varying from 1.69 GHz to 2.33 GHz frequency band under 1.5 V supply. A set of varactors were in assistance to achieve the coarse and exact tuning simultaneously.

I. INTRODUCTION The tremendous advances in telecommunications that concern both the mobile phone sector, and the local networks, as the satellite positioning led to the proliferation of norms and standards [1]. A challenge of research today is to design devices that are reconfigurables, that is to say, be ordered to switch their characteristics from one standard to another [2]. One of the most critical reconfigurable functions is filtering radio frequency. It requires a broadband agreement in order to adapt the templates associated with different standards. Indeed, the most widely used filters are the ones to surface acoustic wave (SAW). These filters are not tunable, and remain one of the most bulky passive devices of RF frontend [3-4]. Active filtering seems to be of solution for these two constraints [5]. Several ways are possible: resonators offset, recursive and transversal filters, Gm-C filters, etc ... For us, we develop on the Q-enhanced LC bandpass filters. In this paper, we outline the highlight features for designing our bandpass filter in section II. The architecture of the Q-enhanced LC filter is exposed in section III. Section IV presents the simulation results of the RF filter. And finally in section V, conclusions are drawn. II.

Q-ENHANCED LC FILTER DESIGN

A simplified Q-enhanced filter is presented in Fig. 1; RP represents equivalent parallel loss resistance which is dominated by the inductor one.

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The negative conductance represents negative resistance which is used in boosting the quality factor in a lossy LC tank. Using positive feedback is the basic idea of Q-Enhanced LC filter. For canceling losses presented by RP, performed on the lossy inductor, a negative resistance can be added, yielding the “parallel mode” Q-enhanced circuit shown in Fig. 2 [6], [7], where: R p (1 Q 2) R s 0

(1)

2 1 Q0 L 2 Q0

Lp

(2)

where Q0 is the self-tuning quality factor of the inductor. In this approach, the negative resistance has been implemented as a transconductor with positive feedback to cancel losses represented by RP. The effective parallel resistance Req and the effective quality factor Qenh of the LC resonator can be found from:

R eq

Rp

1 gm

1 1 gm R p

Rp

(3)

and

Q enh

R eq X Lp

1 Q 1 gm R p 0

(4)

and can be made arbitrarily high by a suitable choice of gm. At large Q enhancements however, gm must be carefully controlled to avoid oscillation (Qenh Æ∞), and circuit tolerances and temperature coefficients become critical issues, requiring the addition of some form of tuning mechanism into the filter design. In addition, the process of Q enhancement creates a regenerative amplification of circuit noise, lowering the resonator’s dynamic range.

A. Frequency tunning To make a filter for multi-standard systems, varactors are an adequate solution to cover a wide frequency range and thus meet many of the standards. One form of this solution is presented in Fig. 3 [8]. It consists in using, instead of a variable capacitor, storage of four parallel varactors which gates are connected to one terminal.

+ + Gm

Vin

Figure 1. Simplified Q-enhanced LC Filter

B. Q-enhancement A primary method for increasing the Q of non-ideal onchip LC tank is the use of negative resistance, implemented by active devices, as shown in Fig. 4. But this is achieved at a cost of higher power consumption and noise, presented by the active devices.

R

V i

V ds1 V ds 2 i

III.

2 V ds g m V ds

2 gm

1 G

L C

C

Lp

Rp

-2/gm

Rs

Figure 2. Parallel mode Q-enhancement

CV1 CV2

Vctr

CV3

CV4

D0 D1 D2 (5)

Digital Word Figure 3. Schematic of the MOS varactors

FILTER ARCHITECTURE

Figure 5 shows the proposed second order Q-enhanced LC bandpass filter. The basic concept in improving the LC filter is to use an LC resonator and to provide partial compensation for its losses by incorporating a negative conductance generator [10], [11]. The input signal is fed to the filter using an input differential transistor pair M1 and M2. The filter’s center frequency is tuned by changing control voltage Vctr and the digital control word for the equivalent varactors capacitance. Negative resistance due to NMOS transistors helps increasing the Q factor of the filter. The filter’s Q is tuned by changing the tail current of the negative conductance circuit built with two cross-coupled NMOS. The varactors are PMOS capacitors operating in accumulation mode within the tuning range. IV.

Vout

-R

Rp

C

-

The drain, the source and the bulk of the varactor CV1 are related to the control voltage Vctr. For varactors CV2, CV3 and CV4 the drain, the source and the bulk are controlled by a digital control word.

A differential topology is used for our balanced circuit which is better in high frequency operation [9], using a crosscoupled transistor pair M1, 2 (gate inputs are connected to opposite drain Outputs). This topology achieves a positive feedback that compensates losses in the LC tank [8]. The voltage to current ratio indicates the effective negative resistance at the terminals M1, 2 as shown in Fig. 4. The impedance seen between the two terminals Vds1 and Vds2 is expressed by R = -2/gm with the simple analysis given by equation (5):

L

SIMULATION RESULTS

AMS CMOS 0.35μm technology has been used for simulation of the bandpass LC filter. As shown in Fig. 6, 7, 8 and 9, the center frequency of the filter can be tuned over

Y

i Vds1

Vds2

M1

M2

IQ Figure 4. Negative resistance implanted by cross-coupled MOSFETs

a 640 MHz range, from 1690 MHz to 2330 MHz. This result is achieved when using varactors in accumulation mode, where a capacity ranges from 2.67 pF to 5.05 pF. Reasonable selectivity is provided for DECT, DCS 1800, PCS 1900 and UMTS applications, while drawing 4mA current from a 1.5V supply. To obtain such filter characteristics, the control voltage of varactors, Vctr, changes from 0.1 V to 1.5 V, and IQ, the bias current, is varied between 0.5 mA and 4 mA.

VDD L1

+ Vin

M1

M2

R1

R2

Vctr

C1

Vout-

L2

C2

M5

Vout+

M6 VDD

-

IQ

VDD IP

M3

M7

M4

M8

Figure 5. Q-Enhanced LC filter Combination 0, 0,0

Combination 1, 0,0

Com bination 0,0,0

Com bination 1,0,0 v db(8,2)

10

v db(8,2)

5

0

-5

5 Voltage Magnitude (dB)

Voltage Magnitude (dB)

Voltage Magnitude (dB)

Voltage Magnitude (dB)

10

0

-5

-10 -10 1.65

1.70

1.75

1.80

1.85

1.90

1.95

1.75

1.80

Frequency (GHz)

1.90

1.95

2.00

2.05

2.10

Frequency (GHz)

Frequency (GHz)

Frequency (GHz)

Figure 6. Frequency response of the filter for first combination.

Figure 8. Frequency response of the filter for third combination.

Combination 1, 1,0

Combination 1, 1,1

Com bination1,1,0

Com bination 1,1,1

10

v db(8,2)

v db(8,2)

5

0

-5

5

Voltage Magnitude (dB)

Voltage Magnitude (dB)

Voltage Magnitude (dB) Voltage Magnitude (dB)

1.85

0

-5

-10 1.85

1.95

2.05 2.10

2.20

Frequency (GHz)

Frequency (GHz)

1.95

2.05

2.15

2.25

2.35

2.45

Frequency (GHz)

Frequency Frequency(GHz) (GHz)

Figure 7. Frequency response of the filter for second combination.

Figure 9. Frequency response of the filter for fourth combination.

Figure 10 shows frequency responses where the Q-factor varies up 80 to 157 when the current IQ varies from 0.5 up to 4 mA for a constant center frequency.

The three main contributors to the nonlinearity of the filter are the negative conductance generator, the varactor and the input gm stage. The nonlinearity analysis of the circuit demonstrates that the contributions of the negative conductance generator and the varactor are much more pronounced than that of the input stage.

The power consumption of the filter is 2.96 mW at 2.1 GHz. The output power of the filter versus input power is shown in figure 11. The 1-dB compression point at the filter’s input is -33.5 dBm.

All simulations of the filter frequency response are using an open circuit load, while noise figure and compression point are simulated using a 50 Ω termination. A summary of the simulated results and those of another work [3] is given in Table I.

inoi se(mag) onoise(mag)

Voltage Magnitude (dB) Voltage Magnitude (dB)

5

0

-5

-10 2.00

2.05

2.10

2.15

2.20

Frequency (GHz) Figure 10. Frequency response of the filter versus IQ(Q-tunning)

P1dB= -33dBm 10

Output power (dBm)

Output power (dBm)

15

5

0

-5

-10

-15 -55

-50

-45

-40

-35

-30

-25

Input power (dBm)

Input power (dBm) Figure 11. 1-dB compression point

-20

2.5 2.0 1.5 1.0 0.5

Input noise 0.0

0.5

1.0

1.5

2.0

2.5

3.0

Frequency (GHz) Figure 12. Noise spectral density of the filter TABLE I.

PERFORMANCE SUMMARY OF RF BANDPASS FILTER Measured value

Parameter Process Filter order Center frequency Quality factor Average -3dB Bandwidth Peaking passband gain Supply voltage Power Noise figure IP1 dB

This work

[3]

AMS 0.35µm CMOS 2 1.69 GHz-2. 33GHz

0.35 µm CMOS 2 1.93 GHz-2.19 GHz

80-157 10.53MHz-29.25MHz

20-170 53.8 MHz

10.99dB-13.91dB

-

1. 5V 2.96 mW 16.65dB -33.5 dBm

1.3 5.2 mW 26.8 dB -30 dBm

REFERENCES [1]

v db(8,2)

3.0

0.0

V. CONCLUSION In this paper we have presented techniques to make possible on-chip LC filters achieving the range of centre frequency from 1.69GHz to 2.33GHz.They are adapted for filter constraints for cellular standards (DECT, DCS1800, PCS 1900 and UMTS). Results have confirmed Q enhancement with negative resistance and the frequency tuning with PMOS varactor array. The improvement of automatic tuning dealing with the distortion in the filter's frequency response is considered as a future work. The design of the multi-band RF CMOS LC bandpass filter is implemented to demonstrate the proposed techniques and to evaluate the potential of such filters. Broader applicability of the techniques presented, such as negative resistance, or varactors array, can be planned because it is easy to fabricate, to control the wide tuning range. Finally, we expect to apply our design features to other kinds of RF on-chip circuits.

10

Output noise

3.5

Noise spectral density Noise Spectral Density (nV/Hz^) (ηV/√Hz)

There are several ways to characterize the noise performance of RF circuits, including the noise spectral density referred to the input, the spectral density of output noise and the noise factor. From these parameters, we can determine the noise of our circuit, which is about 16.65 dB. Fig. 12 shows the noise spectral density figure of the Q-enhanced filter.

C. Andriesei, L. Goras, F. Temcamani, B. Delacressonière, "Wide Tuning Range Active RF Bandpass Filter with MOS Varactors", Romanian Journal of Information Science and Technology (ROMJIST), Volume 12, pp. 485-495,Number 4, 2009. [2] A.Tasic, W.A. Serdijn, J.R. Long. Adaptive multi-standard circuits and systems for wireless communications. IEEE Magazine On Circuits and Systems, vol. 6, no. 1, pp. 29-37, Quarter, 2006. [3] F. Dügler, E. Sánchez-Sinencio, and J. Silva-Martinez, “A 1.3-V 5 mw fully integrated tunable bandpass filter at 2.1 GHz in 0.35-_mCMOS,” IEEE J. Solid-State Circuits, vol. 38, no. 6, pp. 918–928, Jun. 2003. [4] W.B. Kuhn, D. Nobbe, D. Kelly and A.W. Orsborn, “Dynamic range performance of on-chip RF bandpass filters,” IEEE Transactions on Circuits and Systems-II: Analog and Digital Signal Processing, vol. 50, No. 10, pp. 685–694, October 2003. [5] F. Temcamani, B. Delacressonière, M. Dousti, J-L. Gautier, " Filtrage actif dans les systèmes de communication : atouts et défis", Conférence invitée aux 4èmes Journées Franco-Maghrébines des Microondes et de leurs Applications et à TELECOM 2005, Rabat,, March 2005. [6] Y. P. Tsividis, “Integrated continuous-time filter design,” in Proc. IEEE CICC,pp. 641-647.1993. [7] S. Pipilos and Y. Tsividis, “Design of active RLC integrated filters with application in the GHz range,” in Proc. IEEE ISCAS, pp. 5.6455.648.1994. [8] A.W. Orsborn, “Noise analysis and automatic tuning of Q-enhanced LC bandpass filters,” M.S. thesis, Dept. Electr. Comput. Eng., Kansas State Univ., Manhattan, KS, 2001. [9] W. B. Kuhn, F. W. Stephenson, and A. Elshabini-Riad, “A 200- MHz CMOS Q-enhanced LC bandpass filter,” IEEE J. Solid-State Circuits, vol. 31, no. 8, pp. 1112–1122, Aug. 1996. [10] Y. P. Tsividis, “Integrated continuous-time filter design—An overview,” IEEE J. Solid-State Circuits, vol. 29, pp. 166–167, Mar. 1994. [11] D. Li and Y. Tsividis, “Active LC filters on silicon,” IEE Proc.: Circuits, Devices, Syst., vol. 147, no. 1, pp. 49–56, Feb. 2000.